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The bidirectional dcdc converter with high voltage gain and high efficiency plays an important role in the designing of battery charging systems. In this paper, design and development of a battery charging system utilizing coupled inductor based high gain dcdc converter is presented. The converter uses a clamp capacitor network to recover the leakage energy of a coupled inductor. The converter has inherent softswitching capability during turn ON, which ensures high efficiency at high switching frequency. Design equations to derive value of different passive components are given and a stepwise exclusive design to construct coupled inductor is presented. A 50 kHz, 500 W laboratory prototype has been designed, which can increase the voltage with 10 gain (boost operation) in one direction and can reduce the voltage at (1/10) gain (buck operation) in other direction. The CCCV battery charging algorithm is implemented using generic ARM CortexM4 microcontroller. Extensive experiments have been performed and the experimental results are presented in buck, boost, and battery charging operations.
battery charging, bidirectional DCDC converter, high voltage gain, coupled inductor
Due to the intermittent nature of renewable sources, secondary storage device like batteries have become indispensable in many applications. Batteries are an essential part of many power electronic applications like ACDC microgrids, elevators, electric vehicles, etc. [1, 2]. A major technical challenge arises in interfacing batteries with system voltages due to the difference in the level of voltage ratings between them. Design of power electronic converters with required high stepup or stepdown gain becomes an important aspect [3, 4]. Therefore, the interfacing circuits in such applications require a DCDC converter with bidirectional power flow and high voltage conversion ratio. Batteries have limited chargedischarge life period [5, 6]. It is therefore required to charge them with constantcurrent (CC) or constantvoltage (CV) by knowing the state of charge of the batteries. The control of power electronic converters for photovoltaic applications is another task. Some control strategies using Artificial Neural Networks and Fuzzy logic control have been studied by Fapi et al. [7], Jayaraju and Rao [8]. The performance analysis of different controllers for hybrid energy systems has been shown by Katuril and Gorantla [9, 10].
The efficiency of conventional BDC reduces drastically when operated with high duty cycle to achieved HV conversion ratios [11]. This is because of practical circuit drops like device drops, drop due to equivalent series resistance (ESR) of inductor, capacitor etc. The efficiency of such BDC declines with low or higher duty cycle, thereby limiting its conversion ratio [12]. The application of such BDC is limited in battery charging as it cannot achieve HV conversion ratios efficiently. Several BDCs are proposed to achieve energy efficient HV conversion. They are based on high frequency transformer, coupled inductor, switching capacitor, switching inductor, voltage lift technique, voltage multiplier technique etc. [1316].
In highfrequency transformer based BDC topologies, the turns ratio of the transformer provides the HV conversion [17]. The topologies based on dual active bridge are evolved as higher voltage gain bidirectional DCDC converter configurations. These topologies involve a large number of switches resulting in high switching loss, which can be reduced by using softswitching schemes presented by Yang and Do [18]. However, this leads to complex configurations and control. It is also required to protect the switches against HV spikes resulting from the leakage inductance of the isolation transformer. Snubber circuits are utilized to limit these voltage spikes. A snubber capacitor stores the leakage energy during the turn OFF time of switch. This energy is dissipated in the snubber resistor or retrieved by recycling it. Recycling energy requires an active clamping circuit having additional switches and an extra transformer, which makes the BDC bulky and complex [19]. Nonisolated BDC topologies derived from flyback and forward converters have been researched but their implementation is limited to lowpower applications [20, 21]. BDCs based on a switched capacitor have also been researched by Amjadi and Williamson [22]. These arrangements required large number of switches and capacitors for HV conversion, which makes the circuit complicated. The voltage multiplierbased topology needs several cells with high rating, which results in high voltage stresses on the components. Switched inductor and voltage lift technique is not suitable for high power application as it requires a large number of passive components.
In another approach, for HV conversion using BDC, coupled inductorbased topologies are proposed by Hsieh et al. [23]. These converters have less circuit components as compared to the prior transformer and switched capacitorbased topologies. Bipolar core excitation is also possible in these converters, which make them suitable for highpower and high efficiency applications. However, in these arrangements as well, the presence of leakage energy necessitates the use of a clamp capacitor network for the safe operation of switches. High voltage gain of such configurations can be increased with an intermediate capacitor at the secondary of the coupled inductor. Many topologies of coupled inductor based BDC with clamped capacitor have been studied to minimize the numbers of circuit components. Out of them, the BDC proposed by Shreelakshmi et al. [24] has fewest components, i.e., a coupled inductor, a clamp and an intermediate capacitor, and four power electronics switches. In this topology, the efficiency is improved by recovering the leakage energy in the clamp capacitor. All switches achieve an inherent softswitching during turn ON, which reduces switching losses and increases efficiency.
In summary, high gain dcdc converters are essential for interfacing low voltage source with high voltage dc link in the distributed generation environment [25]. The key objective of the presented study are as follows:
1) To develop a charging circuit to charge the lead acid and/or lithiumion batteries. Accordingly, the constant current (CC) and constant voltage (CV) charging methods are implemented.
2) The real time implementation of the coupled inductor based, soft switched, high gain BDC for battery charging application is presented. The complete design steps of the selected converter are demonstrated.
3) The high voltage gain is achieved through the use of a coupled inductor and an intermediate capacitor. The topology ensures ZVS during turnon for all active switches over the entire battery charging range, which enhances the efficiency.
4) The control circuit of the converter is realized using generic lowcost ARM CortexM4 microcontroller STM32F407VG from ST Microelectronics.
5) The converter is designed at 50 kHz high switching frequency, which makes the size of the converter compact. Also, the use of generalpurpose microcontroller makes the converter cost effective.
The implemented high gain BDC consists of a coupled inductor, four switches, a clamp and an intermediate capacitor. The leakage energy is recovered using the clamp capacitor, which improves efficiency. The coupled inductor and intermediate capacitor are utilized to achieve the high voltage gain. At the secondary of the coupled inductor, the intermediate capacitor is charged in series and discharged in parallel during buck operation. The operation is vice versa during the boost mode. The converter has soft switching capability for all switches.
The circuit schematic of the implemented BDC for the battery charging application is shown in Figure 1. The circuit consists of two capacitors and one coupled inductor. C1, C2 are a clamp capacitor and an intermediate capacitor, respectively. There are four switches (S1S4) along with antiparallel body diodes (D1D4). L1 and L2 are the inductances of the low voltage (LV) and high voltage (HV) windings of the coupled inductor, respectively.
In the presented DCDC converter, the leakage energy in the coupled inductor is recovered using the clamp capacitor C1. Due to the recycling of this leakage energy, the losses are less and efficiency is higher in the BDC. The voltage spikes across the switches are also reduced and the converter becomes more reliable. C2 is used for an intermediate energy storage which helps to increase the voltage conversion ratio. Due to the higher voltage conversion ratio and the ZVS capability of the switches, the presented BDC is a suitable choice for battery charging application. The operation of the presented BDC is explained for boost and buck modes in subsequent sections.
Figure 1. Circuit configuration of implemented high gain bidirectional DCDC converter
2.1 Boost mode of operation
(a)
(b)
Figure 2. Steadystate waveforms in boost mode: (a) gate pulses and voltage across switches; and (b) current through different switches
Figure 3. Steadystate waveform of inputoutput parameters of presented BDC during boost mode: V_{LV}, V_{HV}, I_{LV}, and I_{HV}
The boost mode of operation is divided into five intervals as shown in Figure 2 and Figure 3. Figure 2 shows the steadystate waveforms of the current through and the voltage across all switches. Figure 3 shows the steadystate waveforms of the current through and the voltage across the inductor. During interval I, switch S1 is turned ON. The current flows from the battery to the coupled inductor. Energy is stored in the magnetizing inductance of the coupled inductor and the capacitor C2 through the secondary winding of the coupled inductor and the capacitor C1. The body diode D3 is forward biased due to the induced voltage in the coupled inductor. In Interval II, current through D3 becomes zero. The current through S1 increases linearly. This interval ends when S1 is turned OFF.
During interval III, D_{2} is forward biased due to the energy stored in the leakage inductance and the charge of the clamp capacitor C_{1}. Additionally, during this interval the stored coupled inductor energy is transferred to the load through the inductor L_{2}, the capacitor C_{2}, and the diode D_{4}. The interval ends when the current flowing in D_{2} becomes zero. During interval IV, D_{2} turns OFF and D_{4} continues to conduct. This interval ends when the current through D_{4} becomes zero. In interval V, none of the switches are in conduction.
Interval V ends when S_{1} is turned ON. From the steadystate analysis, the voltage gain equation for boost mode can be written as:
$\frac{V_{H V}}{V_{L V}}=\frac{n+1}{1D}$ (1)
where, V_{HV}, V_{LV} are the high voltage and low voltage side voltages, respectively. n is turns ratio of the coupled inductor. D is the duty cycle of switch S_{1}.
2.2 Buck mode of operation
The buck mode of operation is divided into six different intervals as shown in Figure 4. The voltage and current waveforms of all the switches are observed in Figure 4 whereas the voltage and the current waveforms of the coupled inductor are shown in Figure 5. In interval I, S2 is turned ON, and current flows from C1 to the primary winding of the coupled inductor. In this interval, the body diode of S4 starts to conduct when S4 is triggered. ZVS is observed during turn ON of S4 as shown in Figure 4(b). In interval II, only switch S4 conducts and energy is stored in the coupled inductor through the secondary winding. This interval is similar to the turn ON mode of the conventional buck converter. This interval ends when S4 is turned OFF.
(a)
(b)
Figure 4. Steadystate waveforms in buck mode: (a) gate pulses and voltage across switches; and (b) current through different switches
Figure 5. Steadystate waveform of inputoutput parameters of presented BDC during buck mode: V_{LV}, V_{HV}, I_{LV}, and I_{HV}
During interval III, all switches remain OFF, and the body diodes of S_{3} and S_{1} conduct. This interval ends when S_{3} is turned ON. Since D_{3} is already in conduction, ZVS is observed during turn ON of switch S_{3}. Interval IV starts with the turning ON of switch S_{3}. Energy is transferred from C_{2} to C_{1} through the secondary winding of coupled inductor.
The interval V begins when the current through D_{1} becomes zero. Current through S_{3} continues during interval V. This interval ends when S_{3} is turned OFF. Due to turning OFF of S_{3}, the current through secondary of the coupled inductor forces the body diode of switch S_{4} to turn ON. Interval VI starts with the turning ON of diode D_{4}. This interval ends when S_{4} is triggered. From the steadystate analysis, the voltage gain equation for buck mode can be written as:
$\frac{V_{L V}}{V_{H V}}=\frac{D}{1+n}$ (2)
where, D is the duty cycle of switch S4.
The design of the various circuit components of presented high gain DCDC converter is explained in the next section.
In this section, the design of various circuit components is described. The presented BDC is designed for the ratings given in Table 1. The coupled inductor is designed such that it allows for 10 % current ripple in currents. Additionally, all the capacitors have been designed to limit voltage ripple to 0.1 % of rated voltage.
Table 1. Ratings considered for design of different components of BDC
Parameters 
Values 
Power 
500 W 
V_{HV} 
400 V 
V_{LV} 
40 V 
f_{sw} 
50 kHz 
Voltage Ripple 
0.1 % 
Current Ripple 
10 
3.1 Design of coupled inductor
The design of coupled inductor is essential for the BDC. For CCM operation, turns ratio n and duty cycle D is obtained using Eq. (1) and Eq. (2) for boost mode and buck mode, respectively.
For DCM operation, voltage gain of the presented BDC in boost mode is expressed as:
$\frac{V_{H V}}{V_{L V}}=\frac{1+n}{2}+\frac{\sqrt{(1+n)^{2}+\frac{2 D^{2} T_{S} R_{L V}}{L_{1}}}}{2}$ (3)
where, R_{LV} is the low voltage side load resistance. T_{s} is the switching period. L_{1} is the primary side inductance value of the coupled inductor.
The critical inductance of primary winding of coupled inductor L_{1crit} is obtained by equating Eq. (1) and Eq. (3). This is expressed as:
$\frac{L_{1 c r i t}}{T_{S} R_{H V}}=\frac{(1D)^{2} D}{2(1+n)^{2}}$ (4)
where, R_{HV} is the HV side load resistance. L_{1} is the primary side inductance value of the coupled inductor.
The critical inductance of secondary winding L is obtained using the following equation:
$L_{2 c r i t}=n^{2} L_{1 c r i t}$ (5)
where, L_{1crit} is the primary side critical inductance.
The practical values of inductors (L_{1} and L_{2}) are selected as approximately 3 times the critical values for CCM operation. Accordingly, it is calculated to be 45 µH and 720 µH for L_{1} and L_{2}, respectively.
3.2 Design of capacitors
Designs of different capacitors are given in following subsections.
3.2.1 Design of the capacitor C_{HV}
During interval I of boost mode, C_{HV} supplies the load current and its value is calculated as:
$C_{H V}=\frac{I_{H V} D}{f_{s w}(0.001) V_{H V}}$ (6)
where, I_{HV} is the high voltage side source current. f_{sw} denotes the switching frequency.
By considering the coupled inductor current ripple, the value of capacitor C_{HV} is selected as:
$C_{H V}=\frac{\left(\frac{n+1}{1D}1\right) V_{L V}}{8 L_{m} f_{s w}^{2}(0.001) V_{H V}}$ (7)
where, L_{m} is the magnetizing inductance of coupled inductor.
To minimize voltage ripple, higher value amongst Eq. (6) and Eq. (7) is selected.
3.2.2 Design of the capacitor C_{LV}
The capacitor C_{LV} is utilized to filter out inductor current ripple. Based on the ripple current magnitude, the minimum value of the capacitance is calculated as:
$C_{L V}=\frac{1\left(\frac{D}{1+n}\right) V_{L V}}{8 L_{m} f_{s w}^{2}(0.001) V_{L V}}$ (8)
The capacitor C_{LV} also supplies current during the buck operation. Hence, it is evaluated as:
$C_{H V}=\frac{I_{L V} D(0.05)}{f_{s w}(0.001) V_{L V}}$ (9)
where, I_{LV} is the low voltage side source current.
3.2.3 Design of Capacitor C_{2}
This capacitor is charged in parallel with secondary winding during boost operation and is discharged in series with the secondary of the coupled inductor to obtain high conversion ratio. In buck mode, the opposite operation occurs.
$V_{C 2}=\frac{D(1n)+n}{1D} V_{L V}$ (10)
From Eq. (10), the average voltage across the capacitor can be obtained. For the ripple voltage of 0.1 %, The value of capacitor C_{2}is calculated as:
$C_{2}=\frac{I_{L V} D}{n f_{s w}(0.001) V_{C 2}}$ (11)
3.2.4 Design of the capacitor C_{1}
From Eq. (12), the average value of voltage across C_{1} is obtained for buck and boost operation as:
$V_{C 1}=\frac{D V_{H V}}{1+n}$ (12)
In interval II for boost operation, and in interval IV of buck operation, the capacitor C_{1} is charged and value is calculated as:
$C_{1}=\frac{I_{L V} D(0.05)}{f_{s w}(0.001) V_{C 1}}$ (13)
The calculated values for C_{HV}, C_{LV}, C_{2} and C_{1} are 330 µF, 200 µF, 50 µF and 10 µF, respectively.
3.3 Design of switches
The switches are designed to carry RMS current when they are ON and withstand the average drain to source voltage across the switch when they are OFF. The RMS values of switches is observed during simulation study. The basic design equations are mentioned below.
The RMS value of current through the switch S_{1} is calculated as:
$I_{S 1 r m s}=\frac{n V_{L V} T_{S} D \sqrt{D}}{L_{m} \sqrt{3}}$ (14)
The voltage across the switch S_{1} is calculated as:
$V_{S 1}=\frac{1}{lD} V_{L V}$ (15)
The RMS value of current through the switch S_{2} is calculated as:
$I_{S 2 r m s}=\frac{0.05^{1.5} V_{C 1} T_{S} D \sqrt{D}}{L_{l k} \sqrt{3}}$ (16)
The voltage across the switch S_{2} is calculated as:
$V_{S 2}=\frac{1}{1D} V_{L V}$ (17)
The RMS value of current through the switch S3 is calculated as:
$I_{S 3 r m s}=\frac{V_{L V} T_{S} D \sqrt{D}}{L_{m} \sqrt{3}}$ (18)
The voltage across the switch S_{3} is calculated as:
$V_{S 3}=2 n V_{L V}$ (19)
The RMS value of current through the switch S_{4} is calculated as:
$I_{S 4 r m s}=\frac{V_{L V} T_{S} D \sqrt{D}(1D)}{(n+1) L_{1} \sqrt{3}}$ (20)
The voltage across the switch S_{4} is calculated as:
$V_{S 4}=\frac{n}{1D} V_{L V}$ (21)
The rating of the switches for the BDC is shown in Table 2.
3.4 Construction of coupled inductor
This section presents the construction of the coupled inductor using the values of L_{1}and L_{2} as determined in Section 3.1.
3.4.1 Area product
The amount of energy stored in a coupled inductor is calculated as:
$E_{L}=\frac{1}{2} L_{1} I_{m}^{2}$ (22)
where, I_{m} is the maximum magnetizing current through the coupled inductor.
The current in the primary winding during CCM operation increases linearly during the turn ON time and decreases linearly during the turn OFF time. Hence, the average current through the equation is given by:
$I_{m}=I_{\text {L1avg }}+\frac{1}{2} \Delta I_{L 1}$ (23)
where, I_{L1avg} is the average current through inductor and DI_{L1} is the ripple current through the inductor.
Using Eq. (22), the Area Product of the magnetic core is calculated by the following equation:
$A_{P}=A_{w} A_{C}=\frac{2 E_{L}}{K_{w} K_{c} J B_{m}}$ (24)
where, A_{w}, A_{c}, K_{w}, K_{c}, B_{m} and J are the window area of the core, the crosssectional area of the core, the window space factor, the crest factor, the maximum flux density and the current density in the windings, respectively.
From Eq. (24), the minimum required Area Product for the core is calculated as 11253 mm^{4}.
3.4.2 Selection of core and permeance calculation
The core to be selected should have an area product greater than the one calculated in Eq. (24). The permeance of the core is expressed as:
$\Lambda=\frac{\mu_{0} \mu_{r} A_{C}}{l_{m}+\mu_{r} l_{g}}$ (25)
where, Λ, µ_{r}, l_{m}, l_{g} are the permeance, the relative permeability, the magnetic path length and the air gap length, respectively.
Using Eq. (25), the calculated values of permeance of core is 442 nH /turns^{2}.
3.4.3 Calculation of number of turns
The number of turns can be expressed in terms of permeance and inductance as:
$N=\sqrt{\frac{L}{\Lambda}}$ (26)
where, N is the number of turns of the inductor.
The number of turns for primary and secondary windings are calculated as 10 and 40, respectively.
3.4.4 Wire gauge selection
Selection of wire gauge is dependent on the current density of the material and current flowing through to it. The cross section area of the wire is given as:
$a=\frac{I_{r m s}}{J}$ (27)
where, a is the crosssectional area of the wire, I_{rms} is the RMS current flowing through the wire and J is the current density.
For the designed BDC, the current density of both primary and secondary windings is equal. The practical value of a is approximated to the nearest value in the SWG table. The calculated value for a_{1} and a_{2} are 4.166 mm^{2} and 0.4166 mm^{2}, respectively. Accordingly, the selected wire gauges are the SWG 13 and SWG 22.
3.4.5 Check for window area available
To accommodate the number of turns calculated using Eq. (26) with the wire gauge selected using Eq. (27), enough space should be available in the window to accommodate the turns. It is confirmed using the following inequality:
$A_{w} K_{w}>a_{1} N_{1}+a_{2} N_{2}$ (28)
All calculated parameters of circuit components are shown in Table 2. By considering the derived value of components for presented BDC, the experimental setup is constructed. The operation of the circuit is verified in boost and buck modes. Moreover, the CCCV battery charging operation is experimented using the designed BDC. The experimental results are discussed in the next section.
Table 2. Calculated value of different components
Component 
Values 
Coupled Inductor: 

Duty Ratio (D) 
0.5 
Turns Ratio (n) 
4 
Primary Turns (N_{1}) 
10 
Secondary Turns (N_{2}) 
40 
Primary Inductance (L_{1}) 
45 
Secondary Inductance (L_{2}) 
720 
Window Space Factor (K_{w}) 
0.3 
Crest Factor (K_{c}) 
1.05 
Current Density (J) 
3 x 10^6 
Maximum Flux Density (B_{m}) 
0.8 
Area Product required (A_{p}) 
11253 
Outer Diameter (D_{out}) 
52 
Inner Diameter (D_{in}) 
28 
Magnetic Path Length (l_{m}) 
126 
Air Gap Length (l_{g}) 
0 
CrossSection Area of Core (A_{c}) 
180 
Window Area of Core (A_{w}) 
615 
Relative permeability (μ_{r}) 
245 
Permeance of core (Λ) 
442 
Cross Section Area of primary winding 
SWG 13 
Cross Section Area of secondary winding 
SWG 22 
Capacitors 

Input Capacitance (C_{HV}) 
330 μF, 450 V 
Output Capacitance (C_{LV}) 
200 μF, 160 V 
Clamp Capacitor (C_{1}) 
10 μF, 250 V 
Intermediate Capacitor (C_{2}) 
50 μF, 160 V 
Switches 

S_{1} 
100 V, 15 A 
S_{2} 
100 V, 15 A 
S_{3} 
500 V, 5 A 
S_{4} 
500 V, 5 A 
The experimental setup is constructed according to the block diagram shown in Figure 6(a). At the LV side, a leadacid battery rated 12 V 42 Ah is connected whereas at the HV side, DC load of 320 W and rectified DC voltage source are connected using contactor switches. The BDC consists of SiC MOSFETs (C3M0120090D) as the switches. Hall effectbased DC sensor cards (Nitech make) sense the battery voltage (V_{bat}) and current (I_{bat}). These sensed signals are given to the microcontroller, which generates 50 kHz PWM pulses (G_{1}G_{4}) according to the implemented control system. These pulses are processed though the driver card consisting of a TLP250 optocoupler, which provides optical isolation between the power circuit and control circuit. These isolated pulses drive the SiC switches of the designed BDC. The photograph of the experimental prototype is shown in Figure 6(b).
Figure 6. Experimental setup of high gain bidirectional DCDC converter: (a) block diagram; and (b) photograph of the prototype
The control is implemented using a generalpurpose ARM CortexM4 32bit microcontroller, STM32F407VG. Selected control operates at 168 MHz, and it has required onchip 12 bit analog to digital converter (ADC) and dedicated high speed PWM timers. All the voltage and current values are measured utilizing two analog voltmeters and two digital voltmeters. During the experimentation, current and voltage waveforms are captured using a digital storage oscilloscope.
The experiment results are discussed next in three parts. The first and second part presents the performance of the designed BDC in boost mode and buck mode, respectively. These separate experiments are performed to verify the gain and switch voltage stress during the buck and boost mode of the operation. In the third part, the CCCV battery charging scheme is implemented, and results are discussed.
4.1 Experimentation in boost mode of operation
Figure 7. Steadystate waveform of switch voltage and switch current in boost mode: (a) V_{S1}, I_{S1}; (b) V_{S2}, I_{DS2}; (c) V_{S3}, I_{DS3}; and (d) V_{S4}, I_{S4}
Figure 8. Steadystate waveform of voltage and current in boost mode: (a) primary winding: V_{L1}, I_{L1}; (b) secondary winding: V_{L2}, I_{L1}; (c) low voltage input: V_{LV}, I_{LV}; and (d) high voltage output: V_{HV}, I_{HV}
The experimental results of the boost mode are presented in this section. A 12 V battery is used as a source on the LV side of the BDC. On the HV side, a 320 W rheostat is connected as a load. During boost operation, only switch S_{1} is operated. Current and voltage waveforms of switch S_{1} are shown in Figure 7(a). S_{1} is ON in the interval I and II of boost operation. S_{1} helps in storing energy from battery to coupled inductor through the primary winding. Figure 7(b) shows the current and voltage waveform of switch S_{2}. Current I_{DS2} flows through the antiparallel diode D_{2} of switch S_{2}. The leakage energy of coupled inductor is transferred to C_{1} through D_{2} in interval III of boost operation.
The experimental waveform of the current in diode D_{3} is shown in Figure 7(c). During intervals I and II, D_{3} is forward biased, and energy is transferred from C_{1} to C_{2}. This stored energy is transferred through the diode D_{4} to the output capacitor and load resistance during intervals III, IV and V as shown in Figure 7(d). The voltage and the current waveforms in the primary winding L_{1} and secondary winding L_{2} are shown in Figure 8(a) and Figure 8(b), respectively. The current ratio is found dependent on the external circuit and the amount of energy stored. For the parameters given in Table 3, it is observed that the voltage stress across S_{1} and S_{2} is less than 20.0 V while across S_{3} and S_{4} is around 80.0 V in boost operation.
The experimental waveforms of input current and voltage during boost operation are shown in Figure 8(c). During boost mode, input voltage across the battery is 10.8 V and a current of 2.5 A is drawn from the battery. With 50 % duty ratio, the output voltage is observed as 100.2 V and output current is 0.25 A as shown in Figure 8(d). Voltage gain obtained by the BDC is 10 during boost mode of operation.
4.2 Experimentation in buck mode of operation
The experimental results of buck mode are presented in this section. A singlephase AC variac and diode bridge rectifier is used as a source on the HV side of the BDC. On the LV side, a rheostat is connected to perform buck operation in open loop.
Switch S_{1} is not operated during the buck mode. However, the antiparallel diode D_{1} will conduct during intervals III, IV and V. The voltage and the current waveforms of diode D_{1} is observed in Figure 9(a). Diode D_{1} transfers energy stored in the leakage inductance and the magnetizing inductance of the coupled inductor. Figure 9(b) shows the voltage and current waveforms of switch S_{2}. When S_{2} is turned ON, the energy stored in C_{1} transfers to the primary of the coupled inductor. S_{2} remains ON during interval I and II.
Figure 9. Steadystate waveform of switch voltage and switch current in buck mode: (a) V_{S1}, I_{DS1}; (b) V_{S2}, I_{S2}; (c) V_{S3}, I_{S3}; and (d) V_{S4}, I_{S4}
Current I_{S3} through switch S_{3} and the voltage V_{S3} across switch S_{3} are shown in Figure 9(c). S_{3} is ON during interval IV of the buck operation. Energy transfers from C_{2} to C_{1} and L_{2} through the switch S_{3}. In the buck operation, S_{4} is the main controlling switch, where the duty cycle of S_{4} decides the duty cycle of S_{2} and S_{3}. Switch S_{4} remains ON during interval I and II of the buck operation. Figure 9(d) shows the experimental waveforms of current through S_{4} and voltage across S_{4}.
Figure 10. Steadystate waveform of voltage and current in buck mode: (a) primary winding: V_{L1}, I_{L1}; (b) secondary winding: V_{L2}, I_{L1}; (c) low voltage output: V_{LV}, I_{LV}
Figure 10(a) and Figure 10(b) show the voltage and current waveforms of coupled inductor L_{1} and L_{2} respectively. The coupled inductor was designed for turns ratio n = 4 which is verified by observing the voltage magnitude across primary and secondary of the coupled inductor. Currents through these windings are not proportioned like voltages because it is dependent on the external circuit and the amount of energy stored. The voltage across load resistance V_{LV} and current through load resistance I_{LV} are shown in Figure 10(c). Load current and voltage both are constant and ripple free.
It is observed that the voltage stress across S1 and S2 is around 30 V while across S3 and S4 is 120 V in buck operation. For the input voltage of 140 V, output of 13.4 V is observed for 50% duty ratio. Hence, an approximate gain of 1/10 is achieved experimentally. The buck mode of operation of the BDC is validated by this experiment.
Table 3. Leadacid battery charging parameters
Parameter 
Value 
Input Parameters 

Supply Voltage 
140 V 
Max Input Current 
3 A 
Switching Frequency 
50 kHz 
Output Parameters 

Max Battery Voltage 
14.4 V 
Max Battery Current 
11.25 A 
4.3 Battery charging application
In this section, the BDC is operated to charge a leadacid battery. The selected preset voltage level charging scheme for the sixcell battery is as shown in Figure 11(a). Block diagram of charging voltage regulator is shown in Figure 11(b).
The battery voltage V_{bat} and the battery current I_{bat} is sensed using the DC sensor card. Battery voltage is compared with the reference battery voltage. The error is given to a PI controller to generate a reference battery current which is passed through the saturation block to limit it to less than rated charging current. This reference battery current is compared with actual battery current and the error is given to another PI controller.
According to the error, the PI controller adjusts the duty cycle of switching pulse which is generated by the advanced timer (TIM1) of STM32F407VG. The battery charging parameters for leadacid battery is shown in Table 3.
The programmed values for different phases are given below:
(1) Constant Current Phase: Here, $10 \mathrm{~V} \leq V_{b a t}<14 \mathrm{~V}$ and the constant charging current is 1.5 A.
(2) Constant Voltage Phase: Here, V_{bat} = 14 V
Figure 11. Leadacid battery charging: (a) preset voltage level charging scheme for the sixcell battery; (b) block diagram of charging controller
Figure 12. Leadacid battery charging experiment results:
(a) Graph of experimental results, V_{bat} vs t and I_{bat} vs t; and (b) steadystate waveform of V_{bat} and I_{bat}.
The experimental results for a 12 V 42 Ah leadacid battery are shown in Table 4. In this table, input voltage, input current, output voltage, and output current are noted down with respect to time. The output voltage is battery voltage V_{bat} and the output current is battery charging current I_{bat}. Initially, due to the battery discharging, battery voltage is less than the preset voltage. In this interval, the BDC starts charging through a constant current of 1.5 A. Once the battery voltage reaches the preset voltage of 14 V, the BDC starts charging at a constant voltage of 14.4 V and the current starts reducing. The results of Table 4 are plotted and presented in Figure 12(a). The steadystate waveform of charging voltage and current are depicted in Figure 12(b).
The performance of the BDC is observed for boost, buck, and the battery charging operations. Experimentally, the voltage gain is verified as 1/10 in buck mode and 10 in boost mode of operation. During buck mode, the voltage stress on switches S_{1} and S_{2} is 30 V and voltage stress on switches S_{3} and S_{4} is 120 V. During boost mode, the voltage stress on switches S_{1} and S_{2} is 20 V and voltage stress on switches S_{3} and S_{4} is 80 V.
The CCCV method is used to charge a 12 V 42 Ah leadacid battery. During the charging experiment, the lower voltage battery (12 V) is charged effectively from the high voltage of source of 140 V utilizing the designed BDC. This developed charging circuit can also be used with lithiumion batteries requiring constant current and constant voltage charging. The use of SiC MOSFET and soft switching in the converter makes the losses of the converter least possible. The 50 kHz high switching frequency and SiC MOSFET makes the size of the converter compact. Also, the use of generalpurpose microcontroller makes the converter cost effective.
Table 4. Leadacid battery charging test results
Time (Minute) 
Input Voltage (V) 
Input Current (A) 
Output Voltage (V) 
Output Current (A) 
0 
140 
0.180 
13.00 
1.50 
1 
140 
0.180 
13.15 
1.50 
2 
140 
0.180 
13.20 
1.50 
3 
140 
0.180 
13.27 
1.50 
4 
140 
0.180 
13.30 
1.50 
5 
140 
0.180 
13.34 
1.50 
8 
140 
0.180 
13.37 
1.50 
11 
140 
0.180 
13.38 
1.50 
15 
140 
0.180 
13.40 
1.50 
25 
140 
0.180 
13.47 
1.50 
30 
140 
0.180 
13.56 
1.50 
35 
140 
0.180 
13.62 
1.50 
40 
140 
0.180 
13.69 
1.50 
50 
140 
0.180 
13.81 
1.50 
55 
140 
0.180 
13.87 
1.50 
60 
140 
0.180 
14.00 
1.50 
63 
140 
0.180 
14.00 
1.45 
65 
140 
0.180 
14.00 
1.43 
67 
140 
0.175 
14.00 
1.42 
70 
140 
0.175 
14.00 
1.41 
75 
140 
0.180 
14.00 
1.40 
80 
140 
0.180 
14.00 
1.40 
85 
140 
0.175 
14.00 
1.39 
90 
140 
0.175 
14.00 
1.37 
95 
140 
0.175 
14.00 
1.35 
100 
140 
0.175 
14.00 
1.32 
110 
140 
0.170 
14.00 
1.26 
130 
140 
0.160 
14.00 
1.16 
150 
140 
0.150 
14.00 
1.11 
180 
140 
0.135 
14.00 
1.07 
300 
140 
0.130 
14.00 
1.04 
In this paper, a coupled inductor based BDC having inherent softswitching capability is presented for the application of battery charging. The complete design of circuit parameters including coupled inductor have been presented. The results have verified the capability of the proposed converter in boost, buck, and battery charging modes. The features have been validated by the experimental results using a laboratoryscale 500 W prototype. This battery charger is expected to work well in highpower applications where the storage elements are required to be integrated to existing power and electrical systems such as the microgrid, standalone renewable energy systems, elevators, etc.
a 
Crosssectional area of conductor, mm^{2} 
A_{w} 
Window Area of Core, mm^{2} 
A_{c} 
Crosssectional area of core, mm^{2} 
BDC 
Bidirectional DCDC Converter 
CCM 
Continuous Current Mode 
CCCV 
Constant Current Constant Voltage 
D 
Dimensionless duty ratio 
DCM 
Discontinuous Current Mode 
f_{sw} 
switching frequency, Hz 
I_{bat} 
Battery current, A 
J 
Current density, A/mm^{2} 
L_{crit} 
Critical value of inductance, H 
l_{g} 
Air gap length 
l_{m} 
Magnetic path length 
L_{m} 
Magnetizing inductance, H 
n 
Dimensionless turns ratio 
N_{1} 
Turns of transformer HV side 
N_{2} 
Turns of transformer LV side 
PI 
Proportional Integral 
R_{LV} 
Resistance of low voltage side, W 
SiC 
Silicon Carbide 
RMS 
Root Mean Square 
T_{S} 
Switching time period, s 
V_{bat} 
Battery voltage, V 
ZVS 
Zero Voltage Switching 
Greek symbols 

Λ 
Permeance 
µ_{r} 
Relative permeance 
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